There are several types of power converters which comprise a magnetic storage element, such as a transformer or inductor, a primary switch for coupling power to the magnetic element from an input source, and a rectifying device for coupling power from the storage element to a load when the primary switch is not coupling power thereto. (As is known in the art, the term "power" is the rate at which energy is supplied, coupled, delivered, or utilized. It is often used interchangeably with the term "energy" when describing the general function of power and/or energy converters, and when describing the general function of input energy/power sources. Only the term "power" will therefore be used hereinbelow.) Examples of such converters are: buck converters, boost converters, buck-boost converters, and flyback converters. In operation, the primary switch means is switched in cycles of alternating ON-periods and OFF-periods, coupling power to the storage element during each ON-period. The rectifying device is generally configured to detect the start of each OFF-period, as for example by detecting a reverse voltage generated across the magnetic element and/or itself, and for conducting current built up in the storage element to the load. The rectifying device generally conducts this current until it is interrupted by the primary switch turning on, again reversing the voltages across the magnetic storage element and rectifying device.
As is known in the semiconductor art, many rectifying devices are prone to generating a reverse-recovery current when their voltages are reversed from a positive value (forward conducting, .about.0.6 V) to a negative value (reverse biased). The direction of the reverse-recovery current is opposite to that of the forward conduction current and is much larger in magnitude than the reverse leakage current of the rectifying device. The reverse-recovery current is due to the charge stored in the rectifying .device (e.g., in the semiconductor material) which is needed to support the forward current and continues until the stored charge is removed from the rectifying device. The magnitude of the reverse-recovery current is generally set by conditions of the circuit coupled to the rectifying device. In general, the stored charge in uni-polar devices, such as schottky diodes, is removed more quickly than the stored charge in bipolar devices, such as pn-diodes, and are thus preferred.
The primary switch means and rectifying device of the above-noted converters are typically configured such that a short circuit across either the input power supply or output load develops when a reverse-recovery current is generated in the rectifying device. This, of course, causes power dissipation in the converter and reduces the efficiency of the converter. For this reason, uni-polar rectifying devices are often used to minimize the amount of reverse-recovery current and, hence, the amount of power dissipation.
However, there has been a recent trend in the power conversion arts to replace these rectifying devices with "synchronous rectifiers" to reduce the voltage drop across the rectifying devices, and to thereby reduce the power dissipation. Such a synchronous rectifier generally comprises a transistor device coupled in parallel with a rectifying device. The rectifying device conducts as before, but the transistor device is turned on when the rectifying device is conducting, reducing the voltage drop from .about.0.5 V to .about.0.15 V. The rectifying device is generally integrated with the transistor device as a "body diode" on a single semiconductor chip, and is usually formed by a pn-junction, which is capable of generating a large reverse-recovery current. Unfortunately, the power dissipated by a short circuit in the above-noted converters is often comparable to the power dissipation saved by the lower voltage drop.
The reverse-recovery current and its effects may be more fully appreciated by the following exemplary discussion of the standard buck converter, synchronous rectifying buck converter, and transition resonant buck converter.
A standard buck converter is shown at 10 in FIG. 1. Converter 10 converts input power from an unregulated voltage supply to an output having a regulated voltage which is lower than the input voltage. In operation, transistor Q.sub.1 is periodically switched. When closed, Q.sub.1 charges inductor L.sub.1 and couples power to the load. When open, Q.sub.1 causes inductor L.sub.1 to discharge through diode D.sub.1. The output voltage is regulated by varying the time Q.sub.1 is closed during the switching cycle. If the output voltage of the buck converter is set to a low value, such as 3.3 V to 5.0 V, the efficiency of the converter will be poor due to the relatively large forward voltage drop (0.5 V) across rectifier D.sub.1.
To improve the converter's efficiency, rectifier D.sub.1 may be replaced with a synchronous rectifier switch, which has a forward voltage drop on the order of 0.1 V to 0.2 V. An exemplary synchronous rectifier buck converter is shown at 20 in FIG. 2A. Converter 20 comprises two switching transistors Q.sub.1 and Q.sub.2, each preferably comprising a MOSFET transistor for faster switching and hence low switching losses. In operation, transistors Q.sub.1 and Q.sub.2 are alternately switched such that transistors Q.sub.1 and Q.sub.2 are not switched ON at the same time. If Q.sub.1 and Q.sub.2 were on at the same time, a short circuit would be coupled across the input supply, resulting in possible damage to the supply and/or transistors Q.sub.1 and Q.sub.2. As such, Q.sub.2 is turned off just before Q.sub.1 is turned on, and vice-versa. During the short time intervals when both transistors Q.sub.1 and Q.sub.2 are in an OFF state, the current I.sub.L1 through inductor L.sub.1 completes its path through the body diode of transistor Q.sub.2.
Unfortunately, the body diode of transistor Q.sub.2 is not an ultra-fast recovery rectifier, mainly because it is a parasitic component of the switching transistor and is not optimized for fast recovery. With a slow recovery time, the body diode of Q.sub.2 will conduct a significant current in the reverse direction for an amount of time after the diode voltage reverses from positive to negative. This reverse recovery current is substantially larger than the steady-state reverse leakage current and is mainly determined by the voltage and resistance of the external circuit. The reverse current continues until the minority carriers in the diode material (i.e., semiconductor) have recombined with majority carriers. Accordingly, after Q.sub.1 turns on, the body diode Q.sub.2 conducts current during the recovery time and forms a transient short circuit across the input with transistor Q.sub.1. This increases the power dissipation in converter 20. The currents in transistors Q.sub.1 and Q.sub.2 and the current in inductor L.sub.1 are shown in a timing diagram 25 in FIG. 2B. Also shown in FIG. 2B is the voltage at a node A, which couples transistors Q.sub.1 and Q.sub.2. The current through the body diode has been included into the current for transistor Q.sub.2 and its effects are shown by the large current spikes.
The transition resonant buck converter shown at 30 in FIG. 3A is one prior art approach for addressing the reverse recovery problem of converter 20. In converter 30, transistors Q.sub.1 and Q.sub.2 are retained, but the inductance of inductor L.sub.1 is substantially reduced and a resonant capacitor C.sub.2 is coupled across transistor Q.sub.2. The inductance of L.sub.1 is chosen to give a peak-to-peak inductor ripple current of more than twice the maximum load current, as opposed to 10%-20% for converters 10 and 20. The currents in capacitor C.sub.2, transistors Q.sub.1 and Q.sub.2, and inductor L.sub.1 are shown in a timing diagram 35 in FIG. 3B. Also shown in FIG. 3B is the voltage at a node A, which couples transistors Q.sub.1 and Q.sub.2. In the operation of converter 30, the direction of current flow in inductor L.sub.1 alternates between flowing to capacitor C.sub.1 and the load (positive ampere value) and flowing towards resonant capacitor C.sub.2 and transistors Q.sub.1 and Q.sub.2 (negative ampere value) because of the small inductance of L.sub.1. Transistors Q.sub.1 and Q.sub.2 switch in an alternating fashion as in converter 20, with neither switch being on at the same time. During the ON period of each transistor of converter 30, the direction of current flow in inductor L.sub.1 reverses.
More specifically, during the ON-period of Q.sub.1, the current flow through Q.sub.1 ramps from a negative value to a positive value. The body diode of Q.sub.1 may conduct the initial portion of the negative current. Once becoming fully conductive, the MOSFET device of Q.sub.1 effectively short circuits Q.sub.1 's body diode and, with positive current flow in the latter portion of Q.sub.1 's ON period, eliminates the possibility of the diode generating a reverse-recovery current. When Q.sub.1 is turned off, inductor L.sub.1 draws current from resonant capacitor C.sub.2, discharging capacitor C: and reducing the voltage at node A. At zero volts at node A, the body diode of Q.sub.2 conducts and the MOSFET device of Q.sub.2 is subsequently turned on. During Q.sub.2 's ON-period, the current flow through Q: ramps from a negative value to a positive value. Once becoming fully conductive, the MOSFET device of Q.sub.2 effectively short circuits Q.sub.2 's body diode and, with positive current flow in the latter portion of Q.sub.2 's ON period, eliminates the possibility of the diode generating a reverse-recovery current. When Q.sub.2 is turned off, inductor L.sub.1 couples current to capacitor C.sub.2, charging capacitor C.sub.2 and raising the voltage at node A. When the voltage at node A becomes greater than the input voltage, the body diode of Q.sub.1 conducts and the MOSFET device of Q.sub.1 is subsequently turned on for another switching cycle.
In this way, converter 30 eliminates the transient short circuits across the input supply and thereby reduces switching losses. Converter 30 also reduces switching losses by causing each of transistors Q.sub.1 and Q.sub.2 to close when a substantially zero-voltage condition exists. Additionally, converter 30 reduces electromagnetic interference (EMI). As is known in the art, EMI is generated by rapid changes in current around a loop, as for example caused by current spikes, and by rapid changes in node voltages. Both sources of interference are present in standard buck converter 10 and synchronous converter 20 due to the switching of transistors Q.sub.1 and Q.sub.2. Converter 30 has much lower EMI due to the elimination of current spikes and because resonant capacitor C.sub.2 slows down the rate of change in voltage across the transistors and the rate of change in current around circuit loops.
However, transient resonant buck converter 30 has a number of significant disadvantages associated with the large ripple current in inductor L.sub.1 necessary for converter 30 to function properly. This ripple current is approximately five to ten times larger than the ripple currents of converters 10 and 20. As a first disadvantage, the AC component of inductor L.sub.1 's current flows into output capacitor C.sub.1 and the DC component flows into the load. Output capacitor C.sub.1 must be five to ten times larger than that required for converters 10 and 20 to have the same amount of output ripple voltage, and must be a high-quality type to carry the large ripple current without self-heating. The large capacitor value and high quality requirements mean that capacitor C.sub.1 is physically larger and more expensive than the output capacitor for converters 10 and 20. Also, the larger AC ripple current causes the RMS and peak currents in transistors Q.sub.1 and Q.sub.2 to be larger than the corresponding currents in converters 10 and 20. This increases conduction losses in these transistors, which partly offsets the reduction realized by eliminating the recovery current. Additionally, because of the large AC ripple current, the power conversion efficiency of converter 30 at low load-current levels is poor.
As such, the above-described prior art approaches for minimizing the power and energy dissipation caused by reverse-recovery currents have created other power dissipation factors and higher costs. There is therefore a need in the power conversion art for a means of efficiently reducing the power dissipation losses due to reverse-recovery currents generated by rectifying devices and synchronous rectifying devices.